The present invention relates to a switching power supply apparatus to be used to supply a DC voltage to an electronic apparatus and a controlling method for the same.
In recent years, in addition to lower price and down-sizing prerequisites, high performance and energy saving prerequisites have been strongly demanded for industrial and consumer electronic apparatuses. According to such prerequisites, it is also demanded for a switching power supply apparatus for use to supply a DC voltage to the electronic apparatus to be smaller in size, more stable in output operation, and high efficiency.
Hereafter, a conventional switching power supply apparatus will be described with reference to FIG. 4 showing an equivalent circuit diagram of a conventional switching power supply apparatus.
As shown in FIG. 4, this conventional switching power supply apparatus includes a full bridge converter and a DC power source 51 is connected to the input terminals 52a and 52b of the switching power supply apparatus. In the concrete, a positive electrode of the DC power source 51 is connected to the input terminal 52a, and a negative electrode of the DC power source 51 is connected to the input terminal 52b. The DC power source 51 supplies a predetermined input voltage Vin to the input terminals 52a and 52b.
The conventional switching power supply apparatus comprises a first switching element 53, a second switching element 54 connected serially to the first switching element 53, a third switching element 55, a fourth switching element 56 connected serially to the third switching element 55, and a control circuit 57 used to control the first through fourth switching elements 53 . . . 56.
One end of each of the first and third switching elements 53 and 55 is connected to the input terminal 52a, and one end of each of the second and fourth switching elements 54 and 56 is connected to the input terminal 52b. Each of the first through fourth switching elements 53 . . . 56 includes a parasitic capacitance connected to the corresponding element in parallel. That is, as shown in FIG. 4, the first parasitic capacitance 58 grows across the first switching element 53 in parallel, and the second parasitic capacitance 59 grows across the second switching element 54 in parallel. Similarly, the third parasitic capacitance 60 grows across the third switching element 55 in parallel, and the fourth parasitic capacitance 61 grows across the fourth switching element 56 in parallel.
The control circuit 57 receives a DC output voltage Vout generated at both ends of a smoothing capacitor 66 (to be described later), and controls a ratio of an ON-period and an OFF-period of each of the first through fourth switching elements 53 . . . 56 in order to stabilize the output voltage Vout. In the concrete, the control circuit 57 outputs control signals g1 and g2 to the respective first and second switching elements 53 and 54 in a manner that each of the first and second switching elements 53 and 54 repeats becoming the ON-state and the OFF-state alternately. Furthermore, the control circuit 57 outputs a control signal g3 to the third switching element 55 in a manner that the third switching element 55 repeats the ON-state and the OFF-state at the same timing as that of the second switching element 54. Further, the control circuit 57 outputs the control signal g4 to the fourth switching element 56 in a manner that the fourth switching element 56 repeats becoming ON-state and the OFF-state at the same timing as that of the first switching timing 53.
The conventional switching power supply apparatus is provided with a transformer 62 comprises the above-mentioned full bridge converter in addition to the first through fourth switching elements 53 . . . 56. The transformer 62 comprises a primary winding 62a, and first and second secondary windings 62b and 62c. The primary winding 62a is connected to the primary side of the full bridge converter. That is, one end of the primary winding 62a is connected to a connecting point between the first and second switching elements 53 and 54, and the other end of the primary winding 62a is connected to a connecting point between the third and fourth switching elements 55 and 56. The first and second secondary windings 62b and 62c are connected in series each other. The primary winding 62a, and the first and second secondary windings 62b and 62c are set to a turn ratio of n:1:1.
First and second rectifying diodes 63 and 64 are connected to both ends of the serially-connected first and second secondary windings 62b and 62c, respectively. An inductance element 65 and a smoothing capacitor 66 are serially connected to a connecting point between the first and second rectifying diodes 63 and 64 in that order. A load 68 is connected across both ends of the smoothing capacitor 66 via respective output terminals 67a and 67b. Specifically, an anode of the first rectifying diode 63 is connected to one end of the first secondary winding 62b, and an anode of the second rectifying diode 64 is connected to one end of the second secondary winding 62c. Cathodes of the first and second rectifying diodes 63 and 64 are connected to each other, and at the connecting point between those cathodes is connected one end of the inductance element 65.
The other end of the inductance element 65 is connected to one end of the smoothing capacitor 66. One end of the smoothing capacitor 66 is connected to an output terminal 67a, and the other end is connected to an output terminal 67b. Consequently, the first and second rectifying diodes 63 and 64 rectify an induced voltage generated in the first and second secondary windings 62b and 62c, respectively. The smoothing capacitor 66 smooths a rectified induced voltage, and outputs a smoothed induced voltage to the load 68 via the output terminals 67a and 67b as the output voltage Vout. Since the smoothing capacitor 66 is given an electrostatic capacity enough to smooth the induced voltage from the inductance element 65 and output the smoothed voltage as the output voltage Vout.
Operation of this conventional switching power supply apparatus will be described with reference to FIGS. 4 and 5.
FIG. 5 is a waveform chart showing a pulse waveform of each control signal, and voltage and current waveforms at operating condition of the conventional switching power supply apparatus shown in FIG. 4. In (a) through (j) of FIG. 5, abscissa is graduated with time. In (a) through (j) of FIG. 5, the respective waveforms are drawn with their timing positions (represented by vertical broken lines) in agreement with each other.
In (a) through (d) of FIG. 5, the pulse waveforms indicate the control signals g1 . . . g4, respectively. A voltage V51 in (e) of FIG. 5 indicates the waveform of a voltage applied to the second switching element 54. A voltage V52 in (f) of FIG. 5 indicates the waveform of a voltage applied to the fourth switching element 56. A voltage V5t in (g) of FIG. 5 indicates the waveform of a voltage applied to the primary winding 62a. A current I5t in (h) of FIG. 5 indicates the waveform of a current flowing in the primary winding 62a. A current I51 in (i) of FIG. 5 indicates the waveform of a current flowing in a parallel circuit of the first switching element 53 and the first parasitic capacitance 58.
A current I52 in (j) of FIG. 5 indicates the waveform of a current flowing in a parallel circuit of the second switching element 54 and the second parasitic capacitance 59.
The control circuit 57 outputs the control signals g1 and g4 to the respective first and fourth switching elements 53 and 56 at a timing T50 of FIG. 5, so that the first and fourth switching elements 53 and 56 are in the ON-state concurrently. Consequently, the input voltage Vin is applied to the primary winding 62a, so that the induced voltage (Vin/n) is generated in the first secondary winding 62b. As a result, the first rectifying diode 63 is in a conductive state, and the second rectifying diode 64 is in a non-conductive state. The inductance element 65 receives a voltage (Vin/n-Vout) because the output voltage Vout is already applied to the inductance element 65 from the smoothing capacitor 66.
On the other hand, the current I51 in (i) of FIG. 5 is the sum of an exciting current of the transformer 62 and a converted component current, which is the component such that an exciting current of the inductance element 65 is converted into the current flowing through the primary winding 62a, and passes through the first switching element 53.
Subsequently, the control circuit 57 outputs the control signals g1 and g4 to the respective first and fourth switching elements 53 and 56 at a timing T51, so that the first and fourth switching elements 53 and 56 are in the OFF-state concurrently. At this time, since an exciting energy of the transformer 62 has of a continuous characteristic, a secondary current of the transformer 62 is divided into two paths; one in the first secondary winding 62b and the other in the second secondary winding 62c. Consequently, the first and second rectifying diodes 63 and 64 are in the conductive state, so that the induced voltage of the first and second secondary windings 62b and 62c fall to zero. Thereby, the output voltage Vout is applied to the inductance element 18 reversely.
After this, the control circuit 57 outputs the control signals g2 and g3 to the respective second and third switching elements 54 and 55 at a timing T52, so that the second and third switching elements 54 and 55 are in the ON-state concurrently. Consequently, a voltage (-Vin) is thus applied to the primary winding 62a, so that the induced voltage (Vin/n) is generated in the second secondary winding 62c. As a result, the first rectifying diode 63 is in the non-conductive state, and the second rectifying diode 64 is in the conductive state. The inductance element 65 receives the voltage (Vin/n-Vout) because the output voltage Vout is already applied to the inductance element 65 from the smoothing capacitor 66.
On the other hand, a current is the sum of an exciting current of the transformer 62 and a converted component current, which is the component such that an exciting current of the inductance element 65 is converted into the current flowing through the primary winding 62a, and passes through the second and third switching elements 54 and 55.
Subsequently, the control circuit 57 outputs the control signals g2 and g3 to the respective second and third switching elements 54 and 55 at a timing T53, so that the second and third switching elements 54 and 55 are in the OFF-state concurrently. At this time, since an exciting energy of the transformer 62 has of the continuous characteristic, the secondary current of the transformer 62 is divided into two paths; one in the first secondary winding 62b and the other in the second secondary winding 62c. Consequently, the first and second rectifying diodes 63 and 64 are in the conductive state, respectively, so that the induced voltages of the first and second secondary windings 62b and 62c fall to zero. Thereby, the output voltage Vout is applied to the inductance element 65 reversely.
After this, the control circuit 57 outputs the control signals g1 and g4 to the respective first and fourth switching elements 53 and 56 at a timing T54 just like at the timing T50, so that the first and fourth switching elements 53 and 56 are in the ON-state concurrently. Consequently, the input voltage Vin is applied to the primary winding 62a, thereby the above operations are repeated in the switching power supply apparatus.
Hereafter, description will be made in detail for the ratio of the ON-period and the OFF-period of each of the first through fourth switching elements 53 . . . 56 in the conventional switching power supply apparatus.
As shown in the following equation (11), an ON-period Ton where each of the switching elements 53 . . . 56 is in the ON-state is a constant value. Further, as shown in the following equation (12), an OFF-period Toff where all of the switching elements 53 . . . 56 are in the OFF-state is a constant value. EQU Ton=T51-T50=T53-T52 (11) EQU Toff=T52-T51=T54-T53 (12)
When a reset condition of the inductance element 65 shown in the following equation (13) is satisfied, the operation of the inductance element 65 is stabilized. EQU (Vin/n-Vout).times.Ton=Vout.times.Toff (13)
This equation (13) can be modified to the following equation (15) using an equation (14). EQU .delta.=Ton/(Ton+Toff) (14) EQU V=.delta..times.Vin/n (15)
In the conventional switching power supply apparatus, the aforementioned ratio of the ON-period and the OFF-period is adjusted so that a value of ".delta..times.Vin" is a constant value. Thereby, in the conventional switching power supply apparatus, even if the input voltage Vin was varied, it was possible to stabilize the output voltage Vout as shown in the equations (14) and (15).
Furthermore, in this conventional switching power supply apparatus, the full bridge converter composed of the first through fourth switching elements 53 . . . 56 and the transformer 62 was used, and further the input terminals 52a and 52b of this full bridge converter were connected to the DC power source 51. Thereby, in the conventional switching power supply apparatus, a voltage over the input voltage Vin was not applied to any of the first through fourth switching elements 53 . . . 56. In addition, since a well-balanced current is flown in the switching elements 53 . . . 56, the current stress was dispersed and accordingly, it was easy to use the converter for a switching power supply apparatus of a larger electrical power.
However, in the conventional switching power supply apparatus, it was impossible to suppress both surge voltage and surge current generated when each of the first through fourth switching elements 53 . . . 56 was switched between the ON-state and the OFF-state. Therefore, in the conventional switching power supply apparatus, there occurs problems that the surge current caused a power loss, lowering efficiency, and generating noise.
In the concrete, in this conventional switching power supply apparatus, when the first switching element 53 was in the ON-state, for example, at the timing T50 shown in FIG. 5, the first parasitic capacitance 58 was discharged and the second parasitic capacitance 59 was charged. Consequently, the current I51 contained a spike current, which was a surge current (transient current) as shown in (i) of FIG. 5. When the second switching element 54 was in the ON-state, for example, at the timing T52 shown in FIG. 5, the first parasitic capacitance 58 was charged and the second parasitic capacitance 59 was discharged. Consequently, the current I52 contained the spike current as shown in (j) of FIG. 5.
In the same way, when the third switching element 55 was in the ON-state (for example, at the timing T52), the third parasitic capacitance 60 was discharged and the fourth parasitic capacitance 61 was charged. Consequently, a current flowing in the third switching element 55 contained the spike current. When the fourth switching element 56 was in the ON-state (for example, at the timing T50), the third parasitic capacitance 60 was charged and the fourth parasitic capacitance 61 was discharged. Consequently, a current flowing in the fourth switching element 56 contained the spike current.
On the other hand, in this conventional switching power supply apparatus, when each of the first through fourth switching elements 53 . . . 56 was the OFF-state, the surge voltage was generated by a leak inductance of the transformer 62 and parasitic inductance of lead wires, etc. For example, as shown in (g) of FIG. 5, after the first and fourth switching elements 53 and 56 were in the OFF-state at the timing T51, the surge voltage was generated, so that the voltage V5t to be applied to the primary winding 62a was varied. In the same way, after the second and third switching elements 54 and 55 were in the OFF-state at the timing T53, the surge voltage was generated, so that the voltage V5t to be applied to the primary winding 62a was varied.